Readout method and electronic bandwidth control for a silicon in-plane tuning fork gyroscope

ABSTRACT

Disclosed are methods and a sensor architecture that utilizes the residual quadrature error in a gyroscope to achieve and maintain perfect mode-matching, i.e., ˜0 Hz split between the drive and sense mode frequencies, and to electronically control sensor bandwidth. In a reduced-to-practice embodiment, a 6 mW, 3V CMOS ASIC and control algorithm are interfaced to a mode-matched MEMS tuning fork gyroscope to implement an angular rate sensor with bias drift as low as 0.15°/hr and angle random walk of 0.003°/√hr, which is the lowest recorded to date for a silicon MEMS gyroscope. The system bandwidth can be configured between 0.1 Hz and 1 kHz.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.12/218,087, filed Jul. 11, 2008, now U.S. Pat. No. 8,061,201 entitled“READOUT METHOD AND ELECTRONIC BANDWIDTH CONTROL FOR A SILICON IN-PLANETUNING FORK GYROSCOPE,” which claims priority to U.S. ProvisionalApplication Ser. No. 60/949,600, filed Jul. 13, 2007, the entirety ofwhich are incorporated herein by reference for all purposes.

GOVERNMENT RIGHTS

The present invention was funded by the Defense Advanced ResearchProjects Agency (DARPA) under contract No. W31P4Q-04-1-R001. The U.S.Government has certain rights in this invention.

BACKGROUND

The present invention relates generally to microelectromechanicalsystems (MEMS) devices and operational methods, and more particularly,to the use of a zero rate output of a MEMS in-plane tuning forkgyroscope to electronically control the mechanical bias voltages tocontrol the frequency difference between drive and sense resonant modesof the gyroscope.

Over the last decade, the resolution of silicon vibratorymicrogyroscopes has improved by almost ten times every two years. Theimprovements in noise floor can mainly be attributed to improved highaspect ratio microfabrication processes, better mechanical sensordesign, and improved interfacing of the micromechanical sensor elementwith CMOS circuits. Current research is focused on development ofmicrogyroscopes for automotive and consumer applications. However,existing microgyroscope performance must be improved by an order ofmagnitude if they are to be viable alternatives to fiber-opticgyroscopes. Low-cost, sub-degree per hour bias drift microgyroscopeswill complement μ-gravity accelerometers to enable chip-scalenavigation, and multi-axis motion analysis at micro-scale. In addition,such precision inertial measurement units (IMUs) are essential inmicro-robotics, unmanned aerial/undersea vehicles and GPS-augmentednavigation.

Micromachined Coriolis vibratory gyroscopes are ideal angular ratesensors for automotive applications, unmanned aerial vehicles, imagestabilization in portable electronics and personal heading references,due to their low cost, light weight and small form factor. As MEMSgyroscopes attain inertial grade performance (i.e., sub-degree-per-hourrate resolutions and bias stabilities) the interface electronics thatactuate, sense and control these micromechanical structures are keyelement in determining the over all performance of the micro-gyrosystem.

Micromachined gyroscopes constitute one of the fastest growing segmentsof the microsensor market. The application domain of these devices isquickly expanding from automotive to consumer and personal navigationsystems. Examples include anti-skid and safety systems in cars, andimage stabilization in digital cameras. Conventional MEMS gyroscopes donot meet the sub-degree-per-hour resolution and bias drift requirementsneeded in high precision applications such as inertial measurement unitsfor GPS augmented navigation, robotics, unmanned surveillance vehicles,aircraft and personal heading references.

The majority of automotive and consumer electronics application requirerate-grade performance, while high precision navigation-grade devicesare suitable for inertial measurement units and high-end applications inaerospace and petroleum industry.

A multitude of applications exist in the automotive sector includingnavigation, anti-skid and safety systems, roll-over detection, nextgeneration airbag and anti-lock brake systems. Consumer electronicsapplications include image stabilization in digital cameras, smart userinterfaces in handhelds, gaming, and inertial pointing devices. IMUs areself-contained units that can perform accurate short-term navigation ofa craft/object in the absence of global positioning system (GPS)assisted navigation. An IMU typically uses three accelerometers andthree gyroscopes placed along their respective sensitive axes to gatherinformation about an object's direction and heading. MEMS-based IMUs areincreasingly being used in unmanned aerial/undersea vehicles fornavigation and guidance. Since these remotely operated unmannedaerial/undersea vehicles experience diverse environments, in terms ofshock, vibration and temperature, periodic calibration andreconfiguration of the IMU components becomes all the more important.Additionally, these are applications where power and area are premium.This calls for the development of smart angular rate sensors.

Vibratory micromachined gyroscopes rely on Coriolis-induced transfer ofenergy between two vibration modes to sense rotation. Micromachinedgyroscopes are increasingly employed in consumer and automotiveapplications, primarily due to their small size and low powerrequirements. However, they are yet to achieve performance levelscomparable to their optical and macro-mechanical counterparts inhigh-precision applications such as space and tactical/inertialnavigation systems.

Conventional MEMS vibratory gyroscopes have yet to achieve inertialgrade performance. The requirements for inertial grade devices are rateresolutions and bias stabilities better than 0.1°/h. To achieve this, avibratory gyroscope must attain very high quality factors (>30,000),large sense capacitances (>1 pF), large mass (>100 μg), and large driveamplitude (>5 μm).

The Brownian motion of the structure represents the fundamentalnoise-limiting component of a vibratory gyroscope. This is generallydiscussed, for example, by Ayazi, F., in “A High Aspect-RatioHigh-Performance Polysilicon Vibrating Ring Gyroscope,” Ph.D.Dissertation, University of Michigan, Ann Arbor (2001), and Ayazi, F.and Najafi, K., in “A HARPSS Polysilicon Vibrating Ring Gyroscope”IEEE/ASME JMEMS, June 2001, pp. 169-179. By equating Brownian motion tothe displacement caused by the Coriolis force, one can derive themechanical noise equivalent rotation (MNEΩ) of the microgyroscope. Thisis expressed as

${{MNE}\;\Omega} = {{\frac{1}{2q_{Drive}} \cdot \sqrt{\frac{4k_{B}T}{\omega_{0}M}}}\sqrt{BW}}$

This equation indicates that the mechanical noise floor varies inverselywith the drive amplitude (q_(Drive)), the square root of the resonantdrive frequency (ω₀), and square root of the effective mass in the sensedirection (M). Matching the resonant frequencies of the sense and thedrive mode improves this resolution by a factor of √{square root over(Q_(Sense))}.

There is a need for improved tuning fork gyroscopes and angular ratesensors employing same that provide for electronic bandwidth control.

BRIEF DESCRIPTION OF THE DRAWINGS

The various features and advantages of the present invention may be morereadily understood with reference to the following detailed descriptiontaken in conjunction with the accompanying drawing figures, wherein likereference numerals designate like structural element, and in which:

FIG. 1 a illustrates an exemplary tuning fork gyroscope;

FIGS. 1 b and 1 c illustrate resonant mode shapes (exaggerated forclarity) of the gyroscope shown in FIG. 1 a;

FIG. 2 a-2 d illustrates a process flow used to fabricate an exemplarygyroscope;

FIG. 3 illustrates an exemplary sensor comprising the tuning forkgyroscope;

FIG. 4 illustrates mode matching characteristics (physical ZRO) for anexemplary gyroscope;

FIG. 5 illustrates mode matching characteristics (known V_(P)) for anexemplary gyroscope;

FIGS. 6 a and 6 b illustrate bandwidth enhancement and mode matching,respectively, attained by an exemplary gyroscope;

FIG. 7 illustrates an exemplary mode matching algorithm;

FIG. 7 a illustrates exemplary timing for the gyroscope using the modematching algorithm;

FIG. 8 shows a schematic of an exemplary electronic bandwidth controlsystem; and

FIG. 9 plots scale factor (SF) for different bandwidths generated byincrementing and decrementing a V_(P) stepper.

DETAILED DESCRIPTION

The present invention may be used in conjunction with a MEMS in-planetuning fork gyroscope such as is disclosed in U.S. Pat. No. 7,043,985,issued May 16, 2006, and assigned to the assignee of the presentinvention. The contents of U.S. Pat. No. 7,043,985 are incorporatedherein by reference in its entirety.

In order to overcome limitations of conventional microgyroscopes,disclosed are techniques that process the zero rate output of amicro-electromechanical systems in-plane tuning fork gyroscope toelectronically control the mechanical bias voltages such that thefrequency difference between drive and sense resonant modes of themicrogyroscope is nulled. A software control algorithm, or mode-matchingalgorithm, is implemented in conjunction with control circuitry,preferably implemented in a CMOS ASIC, that adaptively biases themechanical structure of the tuning fork gyroscope such that the driveand sense mode frequencies are equalized, or implements bandwidthenhancement at the expense of sensor sensitivity. This leverages theinherent high quality factor of the tuning fork microgyroscope andprovides for an electronically reconfigurable “smart” angular ratesensor with superior sensitivity and bias drift.

Mode-matching of the tuning fork gyroscope may be achieved by increasingthe DC polarization voltage (V_(P)) applied to the MEMS structure(gyroscope) until electrostatic spring softening decreases the sensemode frequency so that it is equal to that of the drive mode (˜0 Hzsplit). Despite quadrature nulling, there always exists a finite amountof residual zero rate output. The amplitude of this residual zero rateoutput is maximized when the modes are matched and can therefore be usedas an indicator of sensor sensitivity. This is a fundamental concept onwhich the software control algorithm achieves mode-matching andimplements a reconfigurable system gain.

The phase difference between the drive output and the zero rate outputis used as an indicator of system stability at the mode-matchedcondition. The mode-matching algorithm performs three steps in eachiteration until a maximum in zero rate output level is detected. Thealgorithm (1) sends an interrupt to a timing unit that generates digitalpulses that read in a ΣΔ bit stream, resets a level detector and updatesa counter values, (2) compares the decimated value of the zero rateoutput level from the current iteration with that from the previous one,and (3) outputs a multi-bit control word to a V_(P)-stepper. Once themaximum is detected, the DC polarization voltage is decremented to itsprevious value, which corresponded to peak sensitivity. Finally, atmatched-mode. the distinct 90° phase difference that exists between thedrive signal (0° drive) and the zero rate output is monitored to ensurethat mode-matching has indeed been achieved.

An exemplary mode-matching ASIC includes a level detector,analog-to-digital converter, a bidirectional counter with parallel load,and a digital-to-analog converter for generating the DC polarizationvoltage in discrete steps (V_(P)-stepper). The architecture of thedigital-to-analog converter is preferably chosen so that the minimumvoltage step size is controlled independently.

The mode-matched condition corresponds to maximum rate sensitivity andlowest noise drift. Another aspect is that the mode-matching algorithmand associated hardware may be used to electronically control systembandwidth inexpensively. If an application requires a larger bandwidth,the modes can be mismatched electronically in a controlled fashion toprovide the required bandwidth at the expense of mechanical gain (i.e.,rate resolution). The controlled mismatch is achieved by incrementing ordecrementing the bidirectional counter to yield the desired value of theDC polarization voltage. Electronic bandwidth control can additionallybe achieved by electronic loading of the mechanical quality factor ofthe drive mode. Further bandwidth control may be achieved by acombination of the above two techniques, i.e., controlled mode-splittingand electronic Q-loading.

Disclosed herein is a smart angular rate sensor system that comprises ahigh performance MEMS tuning fork gyroscope interfaced with CMOScircuitry and a dedicated control algorithm running on amicrocontroller. The system allows software control of parameters of thetuning fork gyroscope, thereby allowing for easy reconfiguration andauto-calibration, in the field. This control technique addresses issuesthat cannot be solved by a simple trimming procedure at the time ofmanufacture. As these sensors experience diverse conditions in thefield, it is impossible to recreate actual field conditions at the timeof manufacture.

The interface architecture and CMOS ASIC substantially improves theperformance of a high-Q tuning fork gyroscope through automatic matchingof its resonant modes, yielding a low-cost microgyroscope with biasdrift of 0.2°/hr. This bias drift is two orders of magnitude better thancommercially available MEMS gyroscopes and is the lowest recorded todate for a MEMS gyroscope.

Referring to the drawing figures, FIG. 1 a illustrates an exemplarymatched-mode in-plane solid-mass single-crystal silicon tuning forkgyroscope 10. The tuning fork gyroscope 10 may be fabricated on 50 μm-60μm thick silicon on insulator (SOI) substrate using a simple two-maskbulk-micromachining process. FIG. 1 a is a top view of the tuning forkgyroscope 10 showing a device layer 23 (FIGS. 3 a-3 d).

It is to be understood, that while the exemplary gyroscope 10 describedherein may be fabricated using silicon, other semiconductor materials,such as quartz or polycrystalline silicon, for example, or anelectrically-conductive substrate, for example, may readily be employed.Thus, it is to be understood that the present invention is not limitedto silicon structures. Also, although the insulating layer is preferablyoxide, it is to be understood, that other insulating material, such asnitride, for example, may also be used, depending upon the application.

The exemplary gyroscope 10 is fabricated as a single-crystal structureusing microelectronic processing techniques. The components making upthe gyroscope 10 are fabricated by depositing, patterning and etchinglayers of semiconductor material and insulating layers to create thedesired interconnected and/or coupled components.

The gyroscope 10 comprises a flexible support structure 11 (alsoreferred to as flexures 11 or beams 11) which is semiconductor materialused to support components of the gyroscope 10. First and second offsetproof masses 12 are supported by the support structure 11. Distal andproximal ends of the proof masses 12 each have a plurality of fingers 12a (or projections 12 a) extending outwardly therefrom. First and seconddrive electrode 13 are disposed adjacent to an outward ends of the firstand second proof masses 12 and have a plurality of fingers 13 a(projections 13 a) extending inwardly therefrom. A central driveelectrode 14 is disposed between the proof masses 12. The central driveelectrode 14 has a plurality of fingers 14 a (projections 14 a)extending outwardly from either lateral edge thereof. A plurality ofsense electrode 15 are disposed adjacent lateral edges of the first andsecond proof masses 12. Quadrature nulling electrodes 17 are laterallydisposed adjacent to the first and second drive electrodes 13. Thequadrature nulling electrodes 17 each have a plurality of fingers 17 a(projections 17 a) extending inwardly therefrom.

The respective fingers 12 a of the proof masses 12 and the fingers 13 a,14 a, 17 a of the drive electrodes 13, 14 and quadrature nullingelectrodes 17 lie adjacent to one another and have air gaps betweenthem. The respective pluralities of fingers 12 a, 13 a, 14 a, 17 a areinterdigitated and comprise comb drive electrodes. As will be discussedbelow, the proof masses 12 are suspended in air, and the first, secondand central drive electrodes 13, 14 are supported by an insulating layer22 (FIGS. 3 a-3 d) and a lower or handle layer 21, or substrate 21(FIGS. 3 a-3 d).

First and second anchors 16 are supported by the insulating layer 22 andhandle layer 21 and are laterally disposed relative to the central driveelectrode 14 and the proof masses 12. The anchors 16 provide support fora tuning fork structure corresponding to the proof masses 12 and driveelectrodes 13, 14. The proof masses 12 and flexible support structure 11that connect proof-masses 12 to the anchors 16 are supported by thehandle layer 21 or substrate 21.

The operating principle of the in-plane tuning fork gyroscope 10 isbased upon a conventional mechanical tuning fork's response to rotation.The twin proof-masses 12 are centrally anchored and driven into resonantoscillations along the x-axis using comb-drive electrodes, i.e.,interdigitated fingers 12 a, 13 a, 14 a, 17 a (drive mode). The drivemode is excited using the central drive electrode 14 to ensure that theproof-masses 12 vibrate anti-phase to one another, preventing lock-in tospurious in-plane modes. Coriolis acceleration induced by rotation aboutthe z-axis is sensed capacitively along the y-axis (sense mode). Theoperating frequency of the gyroscope 10 is approximately 15 kHz, and theanchors 16 and flexures 11 are designed to provide high mechanicalquality factors by minimizing support loss and thermoelastic damping.The fully symmetric and differential Coriolis detection mechanicallyrejects any linear vibration or acceleration as common mode.

A major incentive in using this symmetric tuning fork architecture, asopposed to a conventional frame-design or single-mass-design, isdifferential sensing capability. As a result, linear acceleration/shocksignals are rejected as “common mode” without the need for complexelectronics. The spring-like structure design (provided by the flexiblesupport structure 11) makes it possible to drive the solid proof masses12 linearly with displacement amplitudes in the range of 4-6 μm. A highQ in the drive mode is necessary to produce large drive amplitudes usingsmall drive voltages, which is a highly desirable feature required inlow-power CMOS interfaces. A high Q in sense mode substantiallyincreases sensitivity of the gyroscope 10 and lowers the Brownian noisefloor of the gyroscope 10.

The equations below define the mechanical noise floor (MNEΩ) and currentsensitivity of the tuning fork gyroscope 10, where k_(B) is Boltzmann'sconstant, T is the absolute temperature, q_(Drive) is the amplitude ofthe vibration of the proof-mass 12 along the x-axis, M is the effectivemass, ω₀ is the operating frequency of the sensor 30, V_(P) is the DCpolarization voltage applied to the gyroscope 10, d_(SO) is thecapacitive gap associated with the sensor rest capacitance (C_(SO)), andQ_(EFF) is the effective mechanical quality factor.

${{MNE}\;\Omega} = {\frac{1}{2q_{Drive}}\sqrt{\frac{4k_{B}T}{\omega_{o}{MQ}_{EFF}}}\sqrt{BW}}$$I_{SENSOR} = {\frac{2V_{P}C_{SO}Q_{EFF}q_{drive}}{d_{so}}\Omega_{Z}}$

The primary energy loss mechanism that leads to Q degradation are acombination of support loss, thermoelastic damping and surface roughnessprofile of flexures. This is discussed by Z. Hao, et al., in “AnAnalytical Model for Support Loss in Micromachined Beam Resonators withIn-plane Flexural Vibrations,” Sensors and Actuators A, Vol. 109,December 2003, pp. 156-164. The anchors 16 and the support flexures 11are designed to ensure minimal support loss and consequently high Q inthe drive and sense modes by torque cancellation/reduction. DetailedANSYS simulations of the flexures were performed to optimize structuraldimensions and to allow the sense mode to occur 50-100 Hz higher thanthe drive mode. Once the gyroscope 10 is fabricated, the sense mode istuned electrostatically using the tuning electrodes 14 to match thedrive mode frequency. FIGS. 1 b and 1 c show exemplary resonant modeshapes for an exemplary gyroscope 10 as simulated in an ANSYS computersimulation (exaggerated for clarity). The arrows within the proof masses12 indicate direction of motion.

Quadrature error in the tuning fork gyroscope 10 arises due to off-axismotion of the proof-masses 12 owing to fabrication induced devicemisalignments. This results in unwanted zero rate output (ZRO) at theoperating frequency of the gyroscope 10. In the gyroscope 10, dedicatedquadrature nulling electrodes 17 are added at the outer corners of eachproof-mass 12, as shown in FIG. 1 a. The zero rate output is minimizedin a post fabrication trimming step by applying DC voltages at specificquadrature nulling electrodes 17, which properly aligns the tuning forkgyroscope 10.

This alignment involves application of appropriate rotational torques oneach proof-mass 12, thereby aligning them and decreasing the magnitudeof the zero rate output. However, the quadrature nulling technique onlyreduces the amount of zero rate output in the gyroscope 10, and does noteliminate it altogether.

FIG. 2 a-2 d illustrates fabrication of the gyroscope 10. Referring toFIG. 2 a-2 d, the gyroscope 10 comprises a lower or handle layer 21 orsubstrate 21 and the top or device layer 23, with an insulating layer 22disposed there between. During fabrication, portions of the lower andupper layers 21, 23 and insulating layer 22 are removed usingmicroelectronic fabrication processes to form the gyroscope 10.

More particularly, FIG. 2 a-2 d illustrates a process flow used tofabricate the gyroscope 10 shown in FIG. 1 a. Only a portion of thegyroscope 10 is illustrated in FIG. 2 a-2 d. Moving sections of thegyroscope 10 and areas under the comb drives 18 are first released fromthe backside of the wafer by etching a handle silicon layer 21 throughto a buried oxide layer 22 (insulating layer 22) using a well-knownBosch process. The buried oxide layer 22 is then removed in a reactiveion etching (RIE) system and finally a top layer 23 is patterned all theway through, leaving behind a suspended structure whose anchors 16 aresupported by the handle layer or substrate 21 via several support posts25 of the flexible support structure 11. The final etch step involves ashort hydrofluoric acid (HF) etch to release areas under the supportposts 25.

The fabrication process is very simple and precludes the requirement ofany perforations in the proof mass 12, resulting in a larger mass perunit area. The simultaneous elimination of a ground plane under the combdrives 18 prevents excitation of the out of plane modes and detrimentaleffects of levitation. The fabrication of the device may also beperformed with the same properties using buried cavity wafers.

FIG. 3 illustrates an exemplary angle rate sensor 30 comprising thetuning fork gyroscope 10 and electronic control circuitry 31. There arefour main electronic loops including drive oscillator, sense channel,quadrature nulling and trimming, and automatic mode-matching.

Transresistance front-ends are used in the tuning fork gyroscope 10 toperform capacitance-to-voltage conversion. The T-network transimpedanceamplifiers 35, 61 are relatively immune to parasitic capacitance(C_(TOT)) that is significant in a two-chip implementation of the sensor30, for example. Large transimpedance gains may be implemented in anarea and power-efficient manner on-chip by using the T-networktransimpedance amplifiers 35, 61, as shown in FIG. 3. By strategicsizing of the resistor ratio R2/R3, it is possible to obtain largesignal-to-noise ratios, while keeping noise gain to a minimum. Forexample, continuous time CMOS T-network transimpedance amplifiers 35, 61having a minimum detectable capacitance of 0.02 aF/√Hz and dynamic rangeof 104 dB may be used to perform capacitance-to-voltage conversion alongboth the drive and sense axes.

The drive mode of the gyroscope is excited into mechanical resonanceusing the drive electrodes 13, 14 which are controlled via a drive loop60. The drive loop 60 comprises a high gain T-network transimpedanceamplifier 61 having an input coupled to the first and second driveelectrodes 13 of the gyroscope 10. The T-network transimpedanceamplifier 61 is used in a high gain setting. The phase lock loop 67locks on to the resonant frequency. The phase lock loop-based drive looposcillator 64 provides the required phase shifted signals to sustainelectromechanical drive oscillations and to perform the required signalprocessing operations in the sensor 30. The output of the divide by fourcircuit 65 produces signals that are in-phase (Drv 0°) and inquadrature-phase (Drv 90°) with the velocity of the gyroscopeproof-masses. The in-phase (Drv 0°) signal is applied to the phasedetector circuit 63. The in-phase (Drv 0°) signal is also coupled by wayof a gain control amplifier 66 to the central drive electrode 14 tomaintain electromechanical resonance.

The output signals from each sense transimpedance amplifier 35 areprocessed by a sense channel 41 to produce rate and quadrature outputsignals. When the gyroscope 10 is subject to rotation, a Coriolis signalshows up as an amplitude modulated (AM) signal at the sensor resonantfrequency. The rotation signal is extracted by performing synchronousdemodulation in the sense channel 41 using the output of the phase lockloop 60, which is proportional to the velocity of the proof mass 12. Themultipliers 42, 45 are preferably implemented as Gilbert multipliers 42,45, and the outputs of the multipliers 42, 45 are low-pass filtered toprovide a signal proportional to the input rotation rate. The sensechannel 41 comprises an in-phase multiplier 42 that is coupled through alow pass filter 43 to an in-phase amplifier 44 that produces a rateoutput signal. The in-phase multiplier 42 receives the in-phase (Drv 0°)signal. The sense channel 41 also comprises a quadrature multiplier 445that is coupled through a low pass filer 46 to a quadrature amplifier 47that produces an analog signal that is proportional to the quadratureerror in the gyroscope. The quadrature multiplier 42 receives thequadrature-phase (Drv 90°) signal.

Automatic Mode-Matching

Mode-matching is achieved by increasing the DC polarization voltage(V_(P)) on the MEMS structure until electrostatic spring softeningdecreases the sense mode frequency to become equal to that of the drivemode (˜0 Hz split). The sense frequency is ˜100 Hz higher than the drivefrequency to allow mode-matching despite process variation.

Also, despite quadrature nulling, there exists a finite amount ofresidual zero rate output The amplitude of the residual zero rate outputis maximized when the modes are matched and can therefore be used as anindicator of sensitivity of the sensor 30. Additionally, at mode-matchedcondition, there is a distinct 90° phase shift between the drive outputand ZRO. FIG. 4 illustrates mode matching characteristics (physical ZRO)for an exemplary gyroscope 10. Once matched, synchronous I-Qdemodulation is used to distinguish between the quadrature error and theCoriolis signal. This is a fundamental concept on which the softwarecontrol algorithm 34 achieves mode-matching and provides forreconfigurable system gain.

The software control algorithm 34 provides for electronic tuning of thesense mode resonant frequency to make it equal to the drive moderesonant frequency. This feature differentiates the mode-matched tuningfork gyroscope 10 from other MEMS gyros. More particularly, theelectronic control circuitry 31 includes a computer 32 comprising adecimation and filtering circuitry/software 33 that is coupled to aprocessing engine that implements the software control algorithm 34. Thecomputer 32 and the electronic control circuitry 31 implement automaticmode matching of the tuning fork gyroscope 10.

The MEMS sensor 30 may be interfaced to the electronic control circuitry31, which may comprise a CMOS ASIC, and the control software algorithm34. Alternatively, the CMIOS ASIC can itself contain the softwarealgorithm by using Read-Only-Memory (ROM) and EEPROM. FIG. 7 illustratesan exemplary mode matching algorithm 34 while FIG. 7 a illustratesexemplary timing for the gyroscope 10 using the exemplary mode matchingalgorithm 34. The mechanical imperfections of the sensor 30 areexploited to achieve mode matching automatically.

The exemplary mode matching algorithm 34 is initialized 71 and the zerorate output level is read 72 from the analog-to-digital converter 52.The architecture of the ADC is not restricted to a sigma-delta, but canbe either a successive approximation (SAR) or Flash, as will be evidentto one skilled in the art. The current zero rate output level(ZRO_(CURRENT)) is stored 73. A decision is made 74 whether the currentzero rate output level (ZRO_(CURRENT)) is greater than or equal to theprevious zero rate output level (ZRO_(PREVIOUS)). If the current zerorate output level is less than the previous zero rate output level, thecounter 51 is incremented 75, the polarization voltage (V_(P)) isstepped up 76, and the next zero rate output level is read 72 from theanalog-to-digital converter 52. If the current zero rate output level isgreater than or equal to the previous zero rate output level, thecounter 51 is decremented 77, the polarization voltage (V_(P)) isstepped down 78, and the phase difference (Δφ) is verified 79.

The automatic mode-matching algorithm 34 iteratively increments V_(P)until a maximum in residual ZRO amplitude is detected. The 90° phasedifference is monitored to ensure that mode-matching is achieved. Oncematched, the bandwidth of the sensor 30 can be controlled by varying V,to introduce a controlled frequency separation between the drive andsense resonant modes.

The mode-matching algorithm 34 may be implemented in MATLAB or similarprogramming language and performs three steps in each iteration (FIG. 7)until a maximum in zero rate output level is detected. The algorithm 34(1) sends an interrupt (START) to a timing control circuit 38 thatgenerates digital pulses that read in the bit stream, resets the outputof a level detector 36 and updates the counter values, (2) compares thedecimated value of the zero rate output level from the current iterationwith the value from the previous one, and (3) outputs a 2 bit controlword to the V_(P)-stepper 54. Once the maximum has been detected, V_(P)is decremented to its previous value, which corresponds to peaksensitivity. Exemplary timing is shown in FIG. 7 a. The values of V_(P),and normalized zero rate output levels may be stored in a look-up-table39 (FIG. 8) during mode matching. Finally, at matched mode, the distinct90° phase difference that exists between the drive signal (0° Drv) andthe zero rate output signal is monitored to ensure that mode-matching isachieved. This approach is superior in that it is area and powerefficient, eliminates manual trimming of the gyroscope 10, and providesbandwidth programmability.

T-network transimpedance amplifiers 35 (only one shown) process outputssensed by the respective sense electrodes 15. The zero rate output levelis detected using the level detector 36, which may comprise a 6.5 bitlinear envelope detector that includes a gm-cell followed by a full waverectifier and integrator. In the current embodiment, the DC level may beconverted to a bit-stream using a 1st order ΣΦ modulator 37 read intothe computer 32 via a general purpose interface bus (GPIB) and decimatedusing MATLAB. The integrator in the ΣΦ modulator 37 preferably uses anoise-optimized transconductance amplifier similar to that used in thetransimpedance amplifier 35 and correlated double sampling to mitigateflicker noise and offset. In a reduced-to-practice embodiment, the powerdissipation is 0.9 mW at a clock frequency of 25.6 kHz and measuredSNR_(MAX) for a signal of 100 Hz is 86 dB/Hz.

The bit stream from the ΣΦ modulator 37 is decimated and filtered by thedecimation and filtering circuitry/software 33. Timing for thesigma-delta (ΣΔ) circuit 37 is provided by the timing control circuit38.

The software control algorithm 34 is used to iteratively increment theDC polarization voltage (V_(P)) applied to the MEMS structure (i.e.,anchor 16) until a maximum in residual ZRO amplitude is detected. Thisis accomplished using a V_(P) stepper circuit 54. The V_(P)-stepper 54may include an 8-bit up-down counter 61 and a 6+3 partially segmentedcurrent-steering digital-to-analog (DAC) 52. The up/down counter 51counts up, down or holds the value based on the 2-bit control word. Thealgorithm 34 outputs signals (UP/DN) that increment an up/down counter51. The up/down counter 51 is coupled to an 8-bit digital-to-analogconverter 52, for example, whose output is amplified by a bufferamplifier 53 and applied to the anchor 16 so as to apply and maintain aconstant DC polarization voltage (V_(P)).

The in-phase (Drv 0°) signal is coupled to a phase detector (ΔΦ) circuit68 which also received the zero rate output signal (output from thetransimpedance amplifier). The output of the delta phi (ΔΦ) circuit 68is processed by a phase detector 69 to monitor the phase differencebetween the drive and sense signals. An XOR phase detector 69 may beused to monitor the phase difference between the drive and sensesignals. Comparators in the phase detector 69 ensure that the output ofthe phase detector 69 is purely indicative of phase change and that anyamplitude variations are ignored.

Experimental Results

FIG. 5 shows results of interfacing the matched-mode tuning forkgyroscope 10 to a mode-matching ASIC. In particular, FIG. 5 illustratesmode matching characteristics (known V_(P)) for an exemplary gyroscope10.

The graph shows that the zero rate output increases as thedigital-to-analog converter 52 is incremented. Once the maximum has beencrossed, the zero rate output level falls and the digital-to-analogconverter 52 decrements to its previous value. The phase relationshipsbetween the various signals at matched-mode are shown in FIGS. 6 a and 6b, where the 90° phase difference can be verified. More particularly,FIGS. 6 a and 6 b illustrate bandwidth enhancement and mode matching,respectively, attained by an exemplary gyroscope 10. The frequencydomain mode shapes before and after mode-matching are shown in FIGS. 6 aand 6 b. The matched mode quality factor for this device was 36,000,which illustrates that quadrature nulling and automatic mode-matchingare achieved without any degradation in the mechanical quality factor ofthe sensor 30.

Once matched electronically, the modes are stable and no continuous timemonitoring or recalibration is necessary. Therefore, automaticmode-matching is implemented only when the bandwidth of the sensor 30needs to be reconfigured or when recalibration is necessary. In testsperformed on a reduced-to-practice sensor 30, the 90° phase difference,which is a sensitive indicator of system stability, showed a drift ofonly 0.09° over a period of 75 hours. The bias stability was measuredusing an Allan variance technique at 25° C. and was ˜0.15°/hr, and thecorresponding angle random walk was 0.003°/√hr. These represent thelowest reported values for any silicon-based micromechanical gyroscopeto date. Once matched, the two resonant modes track each other withtemperature, and mode-matching is maintained precluding the need forcontinuous mode monitoring.

Electronic Bandwidth Control

For a micromachined Coriolis vibratory gyroscope, the frequencyseparation between the two resonant modes (Δf) is a measure of theeffective open-loop sensor bandwidth. At perfect mode-matched operation,the linear full-scale range of a reduced-to-practice sensor 30 is ±20deg/second for the maximum achievable drive-mode amplitude of 4 μm andthe effective sensor bandwidth is ˜4 Hz. Such specifications are idealfor high-precision measurement and calibration functions such as gyrocompassing and platform stabilization. While perfect mode matchingprovides exceptional enhancements in bias stability, it places certainlimitations on dynamic range and sensor bandwidth, which are criticalparameters in other applications such as commercial electronics (gamingconsoles) and vehicle electronic stability control.

One particular advantage of the disclosed architecture is that oncematched, the modes can be mismatched by a known amount in a controlledfashion using the up/down counter 51 to obtain a desired bandwidth. Thisallows for electronic bandwidth control, such as is illustrated withreference to FIGS. 6 a, 8 and 9. FIG. 8 shows a schematic of anexemplary electronic bandwidth control system architecture 80.

Referring to FIG. 8, the change in VP (ΔV_(P)) introduces a specificamount of mismatch between the drive and sense modes is calculated fromthe tuning characteristics of the gyroscope 10, and stored in alook-up-table 39. Since the DAC voltage step is fixed, the number ofiterations the DAC 52 must increment or decrement to achieve the desiredfrequency separation is known. The required number of pulses is sent outby the software 34 to a bidirectional counter 51, and the V_(P) stepper54 increments or decrements until the target V_(P) is set on thegyroscope 10. This automatically sets the bandwidth of the microsensor30 without addition of an extra control loop or added hardware.

The disclosed mode-matching architecture can introduce a controlledmismatch (Δf) between drive and sense resonant frequencies, resulting inan open-loop sensor bandwidth equal to Δf. From the tuningcharacteristics, the corresponding ΔV_(P) is obtained, and themode-matching algorithm 34 automatically increments or decrements thecounter 51 to the desired value of V_(P), setting the requiredbandwidth. The resultant decrease in scale factor may be electronicallycompensated if necessary. FIG. 9 plots scale factor (SF) for differentbandwidths generated by incrementing and decrementing the V_(P) stepper54. Tests of a reduced-to-practice sensor 30 indicate that varying V_(P)by 700 mV increases the sensor bandwidth from 1 to ˜10 Hz.

In general terms, the disclosed sensor 30 comprises a gyroscope 10, suchas a microelectromechanical systems tuning fork gyroscope 10 having aplurality of proof masses 12, that is driven into drive mode resonanceby drive electrodes 13 and exhibit sense mode resonance induced byrotation and residual quadrature error sensed by sense electrodes 15.Control circuitry 31 is coupled to the drive electrodes 13 and senseelectrodes 15 that processes the residual quadrature error to generate apolarization voltage applied to the gyroscope 10 to minimizing thefrequency difference between drive and sense resonant modes. The controlcircuitry 31 may be configured to process the residual quadrature errorto apply a polarization voltage to the gyroscope 10 that separates thedrive mode resonance from the sense mode resonance to controloperational bandwidth of the gyroscope 10. The control circuitry 31 maybe further configured to adjusts the gain of the gyroscope 10 inproportion to the change in bandwidth of the gyroscope 10 to control thesensitivity of the gyroscope 10.

Thus, a sub-degree per hour noise floor and bias drift sensor 30comprising a mode-matched tuning fork gyroscope 10 and its associatedcontrol circuitry has been disclosed. Performance benefits are achievedby operating the tuning fork gyroscope 10 in mode-matched condition(i.e., 0 Hz frequency separation between drive and sense resonantmodes). Electronic bandwidth control and automatic mode-matching of thetuning fork gyroscope 10 provide for dynamic configuration of parametersof the tuning fork gyroscope 10. This makes the tuning fork gyroscope 10and sensor 30 ideal for high precision, low bandwidth applicationsincluding gyro compassing, and tactical grade systems requiringbandwidths on the order of 1 to 10 Hz.

Thus, improved tuning fork gyroscopes and angular rate sensors havingelectronic bandwidth control have been disclosed. It is to be understoodthat the above-described embodiments are merely illustrative of some ofthe many specific embodiments that represent applications of theprinciples of the present invention. Clearly, numerous and otherarrangements can be readily devised by those skilled in the art withoutdeparting from the scope of the invention.

The invention claimed is:
 1. A mode matching method for use with amicroelectromechanical systems gyroscope, comprising: determining acurrent zero rate output level of the gyroscope; storing the currentzero rate output level; determining if the current zero rate outputlevel is greater than or equal to a previous zero rate output level; ifthe current zero rate output level is less than the previous zero rateoutput level, incrementing a counter, increasing polarization voltageapplied to the gyroscope, and determining the zero rate output level ofthe gyroscope; if the current zero rate output level is greater than orequal to the previous zero rate output level, decreasing thepolarization voltage applied to the gyroscope, and verifying a phasedifference (.DELTA. .phi.) between drive and sense signals applied toand derived from the gyroscope; iteratively incrementing thepolarization voltage until a maximum in residual zero rate outputamplitude level is detected; monitoring a 90.degree, phase difference toensure that mode-matching is achieved; and once mode matching isachieved, controlling a bandwidth of the gyroscope by varying thepolarization voltage to introduce a controlled frequency separationbetween drive and sense resonant modes of the gyroscope.